High data rate CDMA wireless communication system using variable sized channel codes

ABSTRACT

A method and apparatus for high rate CDMA wireless communication is described. Variable data rates are generated using a set of different encoder, interleaver, and symbol repetition configurations. An encoder associated with each rate generates a variable number of symbols during each frame period. This variable number of symbols is repeated as necessary to form a constant number of symbols equal to a fixed number of symbols that can be then repeated a fixed number of repetitions before transmission. Where the constant number of symbols is not an integer multiple of the variable number of symbols for a particular rate, a subset of the variable number of symbols is repeated to fill in the remaining symbols necessary to equal the constant number of symbols.

This application is a divisional of application Ser. No. 08/856,428,which is a continuation in part of application Ser. No. 08/660,438entitled “REDUCED PEAK-TO-AVERAGE TRANSMIT POWER HIGH DATA RATE CDMAWIRELESS COMMUNICATION SYSTEM” filed Jun. 7, 1996, now U.S. Pat. No.5,926,500, issued Jul. 20, 1999 to Joseph P. Odenwalder, and assigned tothe assignee of the present invention.

BACKGROUND

I. Field

The present invention relates to communications. More particularly, thepresent invention relates to a novel and improved method and apparatusfor high data rate CDMA wireless communication.

II. Description of the Related Art

Wireless communication systems including cellular, satellite and pointto point communication systems use a wireless link comprised of amodulated radio frequency (RF) signal to transmit data between twosystems. The use of a wireless link is desirable for a variety ofreasons including increased mobility and reduced infrastructurerequirements when compared to wire line communication systems. Onedrawback of using a wireless link is the limited amount of communicationcapacity that results from the limited amount of available RF bandwidth.This limited communication capacity is in contrast to wire basedcommunication systems where additional capacity can be added byinstalling additional wire line connections.

Recognizing the limited nature of RF bandwidth, various signalprocessing techniques have been developed for increasing the efficiencywith which wireless communication systems utilize the available RFbandwidth. One widely accepted example of such a bandwidth efficientsignal processing technique is the IS-95 over the air interface standardand its derivatives such as IS-95-A and ANSI J-STD-008 (referred tohereafter collectively as the IS-95 standard) promulgated by thetelecommunication industry association (TIA) and used primarily withincellular telecommunications systems. The IS-95 standard incorporatescode division multiple access (CDMA) signal modulation techniques toconduct multiple communications simultaneously over the same RFbandwidth. When combined with comprehensive power control, conductingmultiple communications over the same bandwidth increases the totalnumber of calls and other communications that can be conducted in awireless communication system by, among other things, increasing thefrequency reuse in comparison to other wireless telecommunicationtechnologies. The use of CDMA techniques in a multiple accesscommunication system is disclosed in U.S. Pat. No. 4,901,307, entitled“SPREAD SPECTRUM COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIALREPEATERS”, and U.S. Pat. No. 5,103,459, entitled “SYSTEM AND METHOD FORGENERATING SIGNAL WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM”, bothof which are assigned to the assignee of the present invention andincorporated by reference herein.

FIG. 1 provides a highly simplified illustration of a cellular telephonesystem configured in accordance with the use of the IS-95 standard.During operation, a set of subscriber units 10 a-d conduct wirelesscommunication by establishing one or more RF interfaces with one or morebase stations 12 a-d using CDMA modulated RF signals. Each RF interfacebetween a base station 12 and a subscriber unit 10 is comprised of aforward link signal transmitted from the base station 12, and a reverselink signal transmitted from the subscriber unit. Using these RFinterfaces, a communication with another user is generally conducted byway of mobile telephone switching office (MTSO) 14 and public switchtelephone network (PSTN) 16. The links between base stations 12, MTSO 14and PSTN 16 are usually formed via wire line connections, although theuse of additional RF or microwave links is also known.

In accordance with the IS-95 standard each subscriber unit 10 transmitsuser data via a single channel, non-coherent, reverse link signal at amaximum data rate of 9.6 or 14.4 kbits/sec depending on which rate setfrom a set of rate sets is selected. A non-coherent link is one in whichphase information is not utilized by the received system. A coherentlink is one in which the receiver exploits knowledge of the carriersignals phase during processing. The phase information typically takesthe form of a pilot signal, but can also be estimated from the datatransmitted. The IS-95 standard calls for a set of sixty four Walshcodes, each comprised of sixty four chips, to be used for the forwardlink.

The use of a single channel, non-coherent, reverse link signal having amaximum data rate of 9.6 of 14.4 kbits/sec as specified by IS-95 is wellsuited for a wireless cellular telephone system in which the typicalcommunication involves the transmission of digitized voice or lower ratedigital data such a facsimile. A non-coherent reverse link was selectedbecause, in a system in which up to 80 subscriber units 10 maycommunicate with a base station 12 for each 1.2288 MHz of bandwidthallocated, providing the necessary pilot data in the transmission fromeach subscriber unit 10 would substantially increase the degree to whicha set of subscriber units 10 interfere with one another. Also, at datarates of 9.6 or 14.4 kbits/sec, the ratio of the transmit power of anypilot data to the user data would be significant, and therefore alsoincrease intersubscriber unit interference. The use of a single channelreverse link signal was chosen because engaging in only one type ofcommunication at a time is consistent with the use of wirelinetelephones, the paradigm on which current wireless cellularcommunications is based. Also, the complexity of processing a singlechannel is less than that associated with processing multiple channels.

As digital communications progress, the demand for wireless transmissionof data for applications such as interactive file browsing and videoteleconferencing is anticipated to increase substantially. This increasewill transform the way in which wireless communications systems areused, and the conditions under which the associated RF interfaces areconducted. In particular, data will be transmitted at higher maximumrates and with a greater variety of possible rates. Also, more reliabletransmission may become necessary as errors in the transmission of dataare less tolerable than errors in the transmission of audio information.Additionally, the increased number of data types will create a need totransmit multiple types of data simultaneously. For example, it may benecessary to exchange a data file while maintaining an audio or videointerface. Also, as the rate of transmission from a subscriber unitincreases the number of subscriber units 10 communicating with a basestation 12 per amount of RF bandwidth will decrease, as the higher datatransmission rates will cause the data processing capacity of the basestation to be reached with fewer subscriber units 10. In some instances,the current IS-95 reverse link may not be ideally suited for all thesechanges. Therefore, the present invention is related to providing ahigher data rate, bandwidth efficient, CDMA interface over whichmultiple types of communication can be performed.

SUMMARY

A novel and improved method and apparatus for high rate CDMA wirelesscommunication is described. In accordance with one embodiment of theinvention, a set of individually gain adjusted subscriber channels areformed via the use of a set of orthogonal subchannel codes having asmall number of PN spreading chips per orthogonal waveform period. Datato be transmitted via one of the transmit channels is low code rateerror correction encoded and sequence repeated before being modulatedwith one of the subchannel codes, gain adjusted, and summed with datamodulated using the other subchannel codes. The resulting summed data ismodulated using a user long code and a pseudorandom spreading code (PNcode) and upconverted for transmission. The use of the short orthogonalcodes provides interference suppression while still allowing extensiveerror correction coding and repetition for time diversity to overcomethe Raleigh fading commonly experienced in terrestrial wireless systems.In the exemplary embodiment of the invention provided, the set ofsub-channel codes are comprised of four Walsh codes, each orthogonal tothe remaining set and four chips in duration. The use of a small number(e.g. four) sub-channels is preferred as it allows shorter orthogonalcodes to be used, however, the use of a greater number of channels andtherefore longer codes is consistent with the invention. In anotherembodiment of the invention the length, or number of chips, in eachchannel code is different to further reduced the peak-to-averagetransmit power.

In a preferred exemplary embodiment of the invention, pilot data istransmitted via a first one of the transmit channels and power controldata transmitted via a second transmit channel. The remaining twotransmit channels are used for transmitting non-specified digital dataincluding user data or signaling data, or both. In an exemplaryembodiment, one of the two non-specified transmit channels is configuredfor BPSK modulation and transmission over the quadrature channel.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, objects, and advantages of the present invention willbecome more apparent from the detailed description set forth below whentaken in conjunction with the drawings in which like referencecharacters identify correspondingly throughout and wherein:

FIG. 1 is a block diagram of cellular telephone system;

FIG. 2 is a block diagram of a subscriber unit and base stationconfigured in accordance with the exemplary embodiment of the invention;

FIG. 3 is a block diagram of a BPSK channel encoder and a QPSK channelencoder configured in accordance with the exemplary embodiment of theinvention;

FIG. 4 is a block diagram of a transmit signal processing systemconfigured in accordance with the exemplary embodiment of the invention;

FIG. 5 is a block diagram of a receive processing system configured inaccordance with the exemplary embodiment of the invention;

FIG. 6 is a block diagram of a finger processing system configured inaccordance with one embodiment of the invention;

FIG. 7 is a block diagram of a BPSK channel decoder and a QPSK channeldecoder configured in accordance with the exemplary embodiment of theinvention; and

FIG. 8 is a block diagram of a transmit signal processing systemconfigured in accordance with a second exemplary embodiment of theinvention;

FIG. 9 is a block diagram of a finger processing system configured inaccordance with one embodiment of the invention;

FIG. 10 is a block diagram of a transmit signal processing systemconfigured in accordance with another embodiment of the invention;

FIG. 11 is a block diagram of the coding performed for the fundamentalchannel when configured in accordance with one embodiment of theinvention;

FIG. 12 is a block diagram of the coding performed for the fundamentalchannel when configured in accordance with one embodiment of theinvention;

FIG. 13 is a block diagram of the coding performed for the supplementalchannel when configured in accordance with one embodiment of theinvention; and

FIG. 14 is a block diagram of the coding performed for the controlchannel when configured in accordance with one embodiment of theinvention.

DETAILED DESCRIPTION

A novel and improved method and apparatus for high rate CDMA wirelesscommunication is described in the context of the reverse linktransmission portion of a cellular telecommunications system. While theinvention is particularly adapted for use within the multipoint-to-pointreverse link transmission of a cellular telephone system, the presentinvention is equally applicable to forward link transmissions. Inaddition, many other wireless communication systems will benefit byincorporation of the invention, including satellite based wirelesscommunication systems, point to point wireless communication systems,and systems transmitting radio frequency signals via the use of co-axialor other broadband cables.

FIG. 2 is a block diagram of receive and transmit systems configured asa subscriber unit 100 and a base station 120 in accordance with oneembodiment of the invention. A first set of data (BPSK data) is receivedby BPSK channel encoder 103, which generates a code symbol streamconfigured for performing BPSK modulation that is received by modulator104. A second set of data (QPSK data) is received by QPSK channelencoder 102, which generates a code symbol stream configured forperforming QPSK modulation that is also received by modulator 104.Modulator 104 also receives power control data and pilot data, which aremodulated along with the BPSK and QPSK encoded data in accordance withcode division multiple access (CDMA) techniques to generate a set ofmodulation symbols received by RF processing system 106. RF processingsystem 106 filters and upconverts the set of modulation symbols to acarrier frequency for transmission to the base station 120 using antenna108. While only one subscriber unit 100 is shown, multiple subscriberunits communicate with base station 120 in the preferred embodiment.

Within base station 120, RF processing system 122 receives thetransmitted RF signals by way of antenna 121 and performs bandpassfiltering, downconversion to baseband, and digitization. Demodulator 124receives the digitized signals and performs demodulation in accordancewith CDMA techniques to produce power control, BPSK, and QPSK softdecision data. BPSK channel decoder 128 decodes the BPSK soft decisiondata received from demodulator 124 to yield a best estimate of the BPSKdata, and QPSK channel decoder 126 decodes the QPSK soft decision datareceived by demodulator 124 to produce a best estimate of the QPSK data.The best estimate of first and second set of data is then available forfurther processing or forwarding to a next destination, and the receivedpower control data used either directly, or after decoding, to adjustthe transmit power of the forward link channel used to transmit data tosubscriber unit 100.

FIG. 3 is a block diagram of BPSK channel encoder 103 and QPSK channelencoder 102 when configured in accordance with the exemplary embodimentof the invention. Within BPSK channel encoder 103 the BPSK data isreceived by CRC check sum generator 130 which generates a check sum foreach 20 ms frame of the first set of data. The frame of data along withthe CRC check sum is received by tail bit generator 132 which appendstail bits comprised of eight logic zeros at the end of each frame toprovide a known state at the end of the decoding process. The frameincluding the code tail bits and CRC check sum is then received byconvolutional encoder 134 which performs, constraint length (K) 9, rate(R) ¼ convolutional encoding thereby generating code symbols at a ratefour times the encoder input rate (E_(R)). In the alternative embodimentof the invention, other encoding rates are performed including rate ½,but the use of rate ¼ is preferred due to its optimalcomplexity-performance characteristics. Block interleaver 136 performsbit interleaving on the code symbols to provide time diversity for morereliable transmission in fast fading environments. The resultinginterleaved symbols are received by variable starting point repeater138, which repeats the interleaved symbol sequence a sufficient numberof times N_(R) to provide a constant rate symbol stream, whichcorresponds to outputting frames having a constant number of symbols.Repeating the symbol sequence also increases the time diversity of thedata to overcome fading. In the exemplary embodiment, the constantnumber of symbols is equal to 6,144 symbols for each frame making thesymbol rate 307.2 kilosymbols per second (ksps). Also, repeater 138 usesa different starting point to begin the repetition for each symbolsequence. When the value of N_(R) necessary to generate 6,144 symbolsper frame is not an integer, the final repetition is only performed fora portion of the symbol sequence. The resulting set of repeated symbolsare received by BPSK mapper 139 which generates a BPSK code symbolstream (BPSK) of +1 and −1 values for performing BPSK modulation. In analternative embodiment of the invention repeater 138 is placed beforeblock interleaver 136 so that block interleaver 136 receives the samenumber of symbols for each frame.

Within QPSK channel encoder 102 the QPSK data is received by CRC checksum generator 140 which generates a check sum for each 20 ms frame. Theframe including the CRC check sum is received by code tail bitsgenerator 142 which appends a set of eight tail bits of logic zeros atthe end of the frame. The frame, now including the code tail bits andCRC check sum, is received by convolutional encoder 144 which performsK=9, R=¼ convolutional encoding thereby generating symbols at a ratefour times the encoder input rate (E_(R)). Block interleaver 146performs bit interleaving on the symbols and the resulting interleavedsymbols are received by variable starting point repeater 148. Variablestarting point repeater 148 repeats the interleaved symbol sequence asufficient number of times N_(R) using a different starting point withinthe symbol sequence for each repetition to generate 12,288 symbols foreach frame making the code symbol rate 614.4 kilosymbols per second(ksps). When N_(R) is not an integer, the final repetition is performedfor only a portion of the symbol sequence. The resulting repeatedsymbols are received by QPSK mapper 149 which generates a QPSK codesymbol stream configured for performing QPSK modulation comprised of anin-phase QPSK code symbol stream of +1 and −1 values (QPSK_(I)), and aquadrature-phase QPSK code symbol stream of +1 and −1 values (QPSK_(Q)).In an alternative embodiment of the invention repeater 148 is placedbefore block interleaver 146 so that block interleaver 146 receives thesame number of symbols for each frame.

FIG. 4 is a block diagram of modulator 104 of FIG. 2 configured inaccordance with the exemplary embodiment of the invention. The BPSKsymbols from BPSK channel encoder 103 are each modulated by Walsh codeW₂ using a multiplier 150 b, and the QPSK_(I) and QPSK_(Q) symbols fromQPSK channel encoder 102 are each modulated with Walsh code W₃ usingmultipliers 150 c and 154 d. The power control data (PC) is modulated byWalsh code W₁ using multiplier 150 a. Gain adjust 152 receives pilotdata (PILOT), which in the preferred embodiment of the invention iscomprised of the logic level associated with positive voltage, andadjusts the amplitude according to a gain adjust factor A₀. The PILOTsignal provides no user data but rather provides phase and amplitudeinformation to the base station so that it can coherently demodulate thedata carried on the remaining sub-channels, and scale the soft-decisionoutput values for combining. Gain adjust 154 adjusts the amplitude ofthe Walsh code W₁ modulated power control data according to gain adjustfactor A₁, and gain adjust 156 adjusts the amplitude of the Walsh codeW₂ modulated BPSK channel data according amplification variable A₂. Gainadjusts 158 a and b adjust the amplitude of the in-phase andquadrature-phase Walsh code W₃ modulated QPSK symbols respectivelyaccording to gain adjust factor A₃. The four Walsh codes used in thepreferred embodiment of the invention are shown in Table I.

TABLE I Modulation Walsh Code Symbols W₀ + + + + W₁ + − + − W₂ + + − −W₃ + − − +

It will be apparent to one skilled in the art that the W₀ code iseffectively no modulation at all, which is consistent with processing ofthe pilot data shown. The power control data is modulated with the W₁code, the BPSK data with the W₂ code, and the QPSK data with the W₃code. Once modulated with the appropriate Walsh code, the pilot, powercontrol data, and BPSK data are transmitted in accordance with BPSKtechniques, and the QPSK data (QPSK_(I) and QPSK_(Q)) in accordance withQPSK techniques as described below. It should also be understood that itis not necessary that every orthogonal channel be used, and that the useof only three of the four Walsh codes where only one user channel isprovided is employed in an alternative embodiment of the invention.

The use of short orthogonal codes generates fewer chips per symbol, andtherefore allows for more extensive coding and repetition when comparedto systems incorporating the use of longer Walsh codes. This moreextensive coding and repetition provides protection against Raleighfading which is a major source of error in terrestrial communicationsystems. The use of other numbers of codes and code lengths isconsistent with the present invention, however, the use of a larger setof longer Walsh codes reduces this enhanced protection against fading.The use of four chip codes is considered optimal because four channelsprovides substantial flexibility for the transmission of various typesof data as illustrated below while also maintaining short code length.

Summer 160 sums the resulting amplitude adjusted modulation symbols fromgain adjusts 152, 154, 156 and 158 a to generate summed modulationsymbols 161. PN spreading codes PN_(I) and PN_(Q) are spread viamultiplication with long code 180 using multipliers 162 a and b. Theresulting pseudorandom code provided by multipliers 162 a and 162 b areused to modulate the summed modulation symbols 161, and gain adjustedquadrature-phase symbols QPSK_(Q) 163, via complex multiplication usingmultipliers 164 a-d and summers 166 a and b. The resulting in-phase termX_(I) and quadrature-phase term X_(Q) are then filtered (filtering notshown), and upconverted to the carrier frequency within RF processingsystem 106 shown in a highly simplified form using multipliers 168 andan in-phase and a quadrature-phase sinusoid. An offset QPSK upconversioncould also be used in an alternative embodiment of the invention. Theresulting in-phase and quadrature-phase upconverted signals are summedusing summer 170 and amplified by master amplifier 172 according tomaster gain adjust A_(M) to generate signal s(t) which is transmitted tobase station 120. In the preferred embodiment of the invention, thesignal is spread and filtered to a 1.2288 MHz bandwidth to remaincompatible with the bandwidth of existing CDMA channels.

By providing multiple orthogonal channels over which data may betransmitted, as well as by using variable rate repeaters that reduce theamount of repeating N_(R) performed in response to high input datarates, the above described method and system of transmit signalprocessing allows a single subscriber unit or other transmit system totransmit data at a variety of data rates. In particular, by decreasingthe rate of repetition N_(R) performed by variable starting pointrepeaters 138 or 148 of FIG. 3, an increasingly higher encoder inputrate E_(R) can be sustained. In an alternative embodiment of theinvention rate ½ convolution encoding is performed with the rate ofrepetition N_(R) increased by two. A set of exemplary encoder ratesE_(R) supported by various rates of repetition N_(R) and encoding ratesR equal to ¼ and ½ for the BPSK channel and the QPSK channel are shownin Tables II and III respectively.

TABLE II BPSK Channel Encoder Encoder Out Out R = ¼ N_(R,R=¼) R = ½N_(R,R=½) E_(R,BPSK) (bits/ (Repetition (bits/ (Repetition Label (bps)frame) Rate, R = ¼) frame) Rate, R = ½) High Rate-72 76,800 6,144  13,072  2 High Rate-64 70,400 5,632  1{fraction (1/11)} 2,816  2{fraction(2/11)} 51,200 4,096  1½ 2,048  3 High Rate-32 38,400 3,072  2 1,536  425,600 2,048  3 1,024  6 RS2-Full Rate 14,400 1,152  5⅓   576  10⅔RS1-Full Rate  9,600   768  8   384  16 NULL   850   68 90{fraction(6/17)}   34 180{fraction (12/17)}

TABLE III QPSK Channel Encoder Encoder Out Out R = ¼ N_(R,R=¼) R = ½N_(R,R=½) E_(R,QPSK) (bits/ (Repetition (bits/ (Repetition Label (bps)frame) Rate, R = ¼) frame) Rate, R = ½) 153,600 12,288  1 6,144  2 HighRate-72  76,800  6,144  2 3,072  4 High Rate-64  70,400  5,632 2{fraction (2/11)} 2,816  4{fraction (4/11)}  51,200  4,096  3 2,048  6High Rate-32  38,400  3,072  4 1,536  8  25,600  2,048  6 1,024  12RS2-Full Rate  14,400  1,152  10⅔   576  21⅓ RS1-Full Rate  9,600   768 16   384  32 NULL    850    68 180{fraction (12/17)}   34 361{fraction(7/17)}

Tables II and III show that by adjusting the number of sequencerepetitions N_(R), a wide variety of data rates can be supportedincluding high data rates, as the encoder input rate E_(R) correspondsto the data transmission rate minus a constant necessary for thetransmission of CRC, code tail bits and any other overhead information.As also shown by tables II and III, QPSK modulation may also be used toincrease the data transmission rate. Rates expected to be used commonlyare provided labels such as “High Rate-72” and “High Rate-32.” Thoserates noted as High Rate-72, High Rate-64, and High Rate-32 have trafficrates of 72, 64 and 32 kbps respectively, plus multiplexed in signalingand other control data with rates of 3.6, 5.2, and 5.2 kbpsrespectively, in the exemplary embodiment of the invention. RatesRS1-Full Rate and RS2-Full Rate correspond to rates used in IS-95compliant communication systems, and therefore are also expected toreceive substantial use for purposes of compatibility. The null rate isthe transmission of a single bit and is used to indicate a frameerasure, which is also part of the IS-95 standard.

The data transmission rate may also be increased by simultaneouslytransmitting data over two or more of the multiple orthogonal channelsperformed either in addition to, or instead of, increasing thetransmission rate via reduction of the repetition rate N_(R). Forexample, a multiplexer (not shown) could split a single data source intoa multiple data sources to be transmitted over multiple datasub-channels. Thus, the total transmit rate can be increased via eithertransmission over a particular channel at higher rates, or multipletransmission performed simultaneously over multiple channels, or both,until the signal processing capability of the receive system is exceededand the error rate becomes unacceptable, or the maximum transmit powerof the of the transmit system power is reached.

Providing multiple channels also enhances flexibility in thetransmission of different types of data. For example, the BPSK channelmay be designated for voice information and the QPSK channel designatedfor transmission of digital data. This embodiment could be moregeneralized by designating one channel for transmission of timesensitive data such as voice at a lower data rate, and designating theother channel for transmission of less time sensitive data such asdigital files. In this embodiment interleaving could be performed inlarger blocks for the less time sensitive data to further increase timediversity. In another embodiment of the invention, the BPSK channelperforms the primary transmission of data, and the QPSK channel performsoverflow transmission. The use of orthogonal Walsh codes eliminates orsubstantially reduces any interference among the set of channelstransmitted from a subscriber unit, and thus minimizes the transmitenergy necessary for their successful reception at the base station.

To increase the processing capability at the receive system, andtherefore increase the extent to which the higher transmissioncapability of the subscriber unit may be utilized, pilot data is alsotransmitted via one of the orthogonal channels. Using the pilot data,coherent processing can be performed at the receive system bydetermining and removing the phase offset of the reverse link signal.Also, the pilot data can be used to optimally weigh multipath signalsreceived with different time delays before being combined in a rakereceiver. Once the phase offset is removed, and the multipath signalsproperly weighted, the multipath signals can be combined decreasing thepower at which the reverse link signal must be received for properprocessing. This decrease in the required receive power allows greatertransmissions rates to be processed successfully, or conversely, theinterference between a set of reverse link signals to be decreased.While some additional transmit power is necessary for the transmissionof the pilot signal, in the context of higher transmission rates theratio of pilot channel power to the total reverse link signal power issubstantially lower than that associated with lower data rate digitalvoice data transmission cellular systems. Thus, within a high data rateCDMA system the E_(b)/N₀ gains achieved by the use of a coherent reverselink outweigh the additional power necessary to transmit pilot data fromeach subscriber unit.

The use of gain adjusts 152-158 as well as master amplifier 172 furtherincreases the degree to which the high transmission capability of theabove described system can be utilized by allowing the transmit systemto adapt to various radio channel conditions, transmission rates, anddata types. In particular, the transmit power of a channel that isnecessary for proper reception may change over time, and with changingconditions, in a manner that is independent of the other orthogonalchannels. For example, during the initial acquisition of the reverselink signal the power of the pilot channel may need to be increased tofacilitate detection and synchronization at the base station. Once thereverse link signal is acquired, however, the necessary transmit powerof the pilot channel would substantially decrease, and would varydepending on various factors including the subscriber units rate ofmovement. Accordingly, the value of the gain adjust factor A₀ would beincreased during signal acquisition, and then reduced during an ongoingcommunication. In another example, when information more tolerable oferror is being transmitted via the forward link, or the environment inwhich the forward link transmission is taking place is not prone to fadeconditions, the gain adjust factor A₁ may be reduced as the need totransmit power control data with a low error rate decreases. In oneembodiment of the invention, whenever power control adjustment is notnecessary the gain adjust factor A₁ is reduced to zero.

In another embodiment of the invention, the ability to gain adjust eachorthogonal channel or the entire reverse link signal is furtherexploited by allowing the base station 120 or other receive system toalter the gain adjust of a channel, or of the entire reverse linksignal, via the use of power control commands transmitted via theforward link signal. In particular, the base station may transmit powercontrol information requesting the transmit power of a particularchannel or the entire reverse link signal be adjusted. This isadvantageous in many instances including when two types of data havingdifferent sensitivity to error, such as digitized voice and digitaldata, are being transmitted via the BPSK and QPSK channels. In thiscase, the base station 120 would establish different target error ratesfor the two associated channels. If the actual error rate of a channelexceeded the target error rate, the base station would instruct thesubscriber unit to reduce the gain adjust of that channel until theactual error rate reached the target error rate. This would eventuallylead to the gain adjust factor of one channel being increased relativeto the other. That is, the gain adjust factor associated with the moreerror sensitive data would be increased relative to the gain adjustfactor associated with the less sensitive data. In other instances, thetransmit power of the entire reverse link may require adjustment due tofade conditions or movement of the subscriber unit 100. In theseinstances, the base station 120 can do so via transmission of a singlepower control command.

Thus, by allowing the gain of the four orthogonal channels to beadjusted independently, as well as in conjunction with one another, thetotal transmit power of the reverse link signal can be kept at theminimum necessary for successful transmission of each data type, whetherit is pilot data, power control data, signaling data, or different typesof user data. Furthermore, successful transmission can be defineddifferently for each data type. Transmitting with the minimum amount ofpower necessary allows the greatest amount of data to be transmitted tothe base station given the finite transmit power capability of asubscriber unit, and also reduces the interfere between subscriberunits. This reduction in interference increases the total communicationcapacity of the entire CDMA wireless cellular system.

The power control channel used in the reverse link signal allows thesubscriber unit to transmit power control information to the basestation at a variety of rates including a rate of 800 power control bitsper second. In the preferred embodiment of the invention, a powercontrol bit instructs the base station to increase or decrease thetransmit power of the forward link traffic channel being used totransmit information to the subscriber unit. While it is generallyuseful to have rapid power control within a CDMA system, it isespecially useful in the context of higher data rate communicationsinvolving data transmission, because digital data is more sensitive toerrors, and the high transmission causes substantial amounts of data tobe lost during even brief fade conditions. Given that a high speedreverse link transmission is likely to be accompanied by a high speedforward link transmission, providing for the rapid transmission of powercontrol over the reverse link further facilitates high speedcommunications within CDMA wireless telecommunications systems.

In an alternative exemplary embodiment of the invention a set of encoderinput rates E_(R) defined by the particular N_(R) are used to transmit aparticular type of data. That is, data may be transmitted at a maximumencoder input rate E_(R) or at a set of lower encoder input rates E_(R),with the associated N_(R) adjusted accordingly. In the preferredimplementation of this embodiment, the maximum rates corresponds to themaximum rates used in IS-95 compliant wireless communication system,referred to above with respect to Tables II and III as RS1-Full Rate andRS2-Full Rate, and each lower rate is approximately one half the nexthigher rate, creating a set of rates comprised of a full rate, a halfrate, a quarter rate, and an eighth rate. The lower data rates arepreferable generated by increasing the symbol repetition rate N_(R) withvalue of N_(R) for rate set one and rate set two in a BPSK channelprovided in Table IV.

TABLE IV RS1 and RS2 Rate Sets in BPSK Channel Encoder Encoder Out Out R= ¼ N_(R,R=¼) R = ½ N_(R,R=½) E_(R,QPSK) (bits/ (Repetition (bits/(Repetition Label (bps) frame) Rate, R = ¼) frame) Rate, R = ½) RS2-FullRate 14,400 1,152  5⅓ 576  10⅔ RS2-Half Rate  7,200   576 10⅔ 288  21⅓RS2-Quarter  3,600   288 21⅓ 144  42⅔ Rate RS2-Eighth  1,900   15240{fraction (8/19)}  76  80{fraction (16/19)} Rate RS1-Full Rate  9,600  768  8 384  16 RS1-Half Rate  4,800   384 16 192  32 RS1-Quarter 2,800   224 27{fraction (3/7)} 112  54{fraction (6/7)} Rate RS1-Eighth 1,600   128 48  64  96 Rate NULL   850   68 90{fraction (6/17)}  34180{fraction (12/17)}

The repetition rates for a QPSK channel is twice that for the BPSKchannel.

In accordance with the exemplary embodiment of the invention, when thedata rate of a frame changes with respect to the previous frame thetransmit power of the frame is adjusted according to the change intransmission rate. That is, when a lower rate frame is transmitted aftera higher rate frame, the transmit power of the transmit channel overwhich the frame is being transmitted is reduced for the lower rate framein proportion to the reduction in rate, and vice versa. For example, ifthe transmit power of a channel during the transmission of a full rateframe is transmit power T, the transmit power during the subsequenttransmission of a half rate frame is transmit power T/2. The reductionis transmit power is preferably performed by reducing the transmit powerfor the entire duration of the frame, but may also be performed byreducing the transmit duty cycle such that some redundant information is“blanked out.” In either case, the transmit power adjustment takes placein combination with a closed loop power control mechanism whereby thetransmit power is further adjusted in response to power control datatransmitted from the base station.

FIG. 5 is a block diagram of RF processing system 122 and demodulator124 of FIG. 2 configured in accordance with the exemplary embodiment ofthe invention. Multipliers 180 a and 180 b downconvert the signalsreceived from antenna 121 with an in-phase sinusoid and a quadraturephase sinusoid producing in-phase receive samples R_(I) andquadrature-phase receive samples R_(Q) receptively. It should beunderstood that RF processing system 122 is shown in a highly simplifiedform, and that the signals are also match filtered and digitized (notshown) in accordance with widely known techniques. Receive samples R_(I)and R_(Q) are then applied to finger demodulators 182 within demodulator124. Each finger demodulator 182 processes an instance of the reverselink signal transmitted by subscriber unit 100, if such an instance isavailable, where each instance of the reverse link signal is generatedvia multipath phenomenon. While three finger demodulators are shown, theuse of alternative numbers of finger processors are consistent with theinvention including the use of a single finger demodulator 182. Eachfinger demodulator 182 produces a set of soft decision data comprised ofpower control data, BPSK data, and QPSK_(I) data and QPSK_(Q) data. Eachset of soft decision data is also time adjusted within the correspondingfinger demodulator 182, although time adjustment could be performedwithin combiner 184 in an alternative embodiment of the invention.Combiner 184 then sums the sets of soft decision data received fromfinger demodulators 182 to yield a single instance of power control,BPSK, QPSK_(I) and QPSK_(Q) soft decision data.

FIG. 6 is block diagram a finger demodulator 182 of FIG. 5 configured inaccordance with the exemplary embodiment of the invention. The R_(I) andR_(Q) receive samples are first time adjusted using time adjust 190 inaccordance with the amount of delay introduced by the transmission pathof the particular instance of the reverse link signal being processed.Long code 200 is mixed with pseudorandom spreading codes PN_(I) andPN_(Q) using multipliers 201, and the complex conjugate of the resultinglong code modulated PN_(I) and PN_(Q) spreading codes are complexmultiplied with the time adjusted R_(I) and R_(Q) receive samples usingmultipliers 202 and summers 204 yielding terms X_(I) and X_(Q). Threeseparate instances of the X_(I) and X_(Q) terms are then demodulatedusing the Walsh codes W₁, W₂ and W₃ respectively, and the resultingWalsh demodulated data is summed over four demodulation chips using 4 to1 summers 212. A fourth instance of the X_(I) and X_(Q) data is summedover four demodulation chips using summers 208, and then filtered usingpilot filters 214. In the preferred embodiment of the invention pilotfilter 214 performs averaging over a series of summations performed bysummers 208, but other filtering techniques will be apparent to oneskilled in the art. The filtered in-phase and quadrature-phase pilotsignals are used to phase rotate and scale the W₁, and W₂ Walsh codedemodulated data in accordance with BPSK modulated data via complexconjugate multiplication using multipliers 216 and adders 217 yieldingsoft decision power control and BPSK data. The W₃ Walsh code modulateddata is phase rotated using the in-phase and quadrature-phase filteredpilot signals in accordance with QPSK modulated data using multipliers218 and adders 220, yielding soft decision QPSK data. The soft decisionpower control data is summed over 384 modulation symbols by 384 to 1summer 222 yielding power control soft decision data. The phase rotatedW₂ Walsh code modulated data, the W₃ Walsh code modulated data, and thepower control soft decision data are then made available for combining.In an alternative embodiment of the invention, encoding and decoding isperformed on the power control data as well.

In addition to providing phase information the pilot may also be usedwithin the receive system to facilitate time tracking. Time tracking isperformed by also processing the received data at one sample time before(early), and one sample time after (late), the present receive samplebeing processed. To determine the time that most closely matches theactual arrival time, the amplitude of the pilot channel at the early andlate sample time can be compared with the amplitude at the presentsample time to determine that which is greatest. If the signal at one ofthe adjacent sample times is greater than that at the present sampletime, the timing can be adjusted so that the best demodulation resultsare obtained.

FIG. 7 is a block diagram of BPSK channel decoder 128 and QPSK channeldecoder 126 (FIG. 2) configured in accordance with the exemplaryembodiment of the invention. BPSK soft decision data from combiner 184(FIG. 5) is received by accumulator 240 which stores the first sequenceof 6,144/N_(R) demodulation symbols in the received frame where N_(R)depends on the transmission rate of the BPSK soft decision data asdescribed above, and adds each subsequent set of 6,144/N_(R) demodulatedsymbols contained in the frame with the corresponding stored accumulatedsymbols. Block deinterleaver 242 deinterleaves the accumulated softdecision data from variable starting point summer 240, and Viterbidecoder 244 decodes the deinterleaved soft decision data to produce harddecision data as well as CRC check sum results. Within QPSK decoder 126QPSK_(I) and QPSK_(Q) soft decision data from combiner 184 (FIG. 5) aredemultiplexed into a single soft decision data stream by demux 246 andthe single soft decision data stream is received by accumulator 248which accumulates every 6,144/N_(R) demodulation symbols where N_(R)depends on the transmission rate of the QPSK data. Block deinterleaver250 deinterleaves the soft decision data from variable starting pointsummer 248, and Viterbi decoder 252 decodes the deinterleaved modulationsymbols to produce hard decision data as well as CRC check sum results.In the alternative exemplary embodiment described above with respect toFIG. 3 in which symbol repetition is performed before interleaving,accumulators 240 and 248 are placed after block deinterleavers 242 and250. In the embodiment of the invention incorporating the use of ratesets, and therefore in which the rate of particular frame is not known,multiple decoders are employed, each operating at a differenttransmission rate, and then the frame associated with the transmissionrate most likely to have been used is selected based on the CRC checksumresults. The use of other error checking methods is consistent with thepractice of the present invention.

FIG. 8 is a block diagram of modulator 104 (FIG. 2) configured in analternative embodiment of the invention in which a single BPSK datachannel is employed. Pilot data is gain adjusted by gain adjust 452 inaccordance with gain adjust factor A₀. Power control data is modulatedwith Walsh code W₁ by multiplier 150 a and gain adjusted by gain adjust454 in accordance with gain adjust factor A₁. The gain adjusted pilotdata and power control data are summed by summer 460 producing summeddata 461. BPSK data is modulated with Walsh code W₂ by multiplier 150 band then gain adjusted using gain adjust 456 in accordance with gainadjust factor A₂.

In-phase pseudo random spreading code (PN_(I)) and quadrature-phasepseudo random spreading code (PN_(Q)) are both modulated with long code480. The resulting long code modulated PN_(I) and PN_(Q) codes arecomplex multiplied with the summed data 461 and the gain adjusted BPSKdata from gain adjust 456 using multipliers 464 a-d and summers 466 a-byielding terms X_(I) and X_(Q). Terms X_(I) and X_(Q) are thenupconverted with in-phase and quadrature-phase sinusoids suingmultipliers 468 and the resulting upconverted signals are summed bysummers 470 respectively, and amplified by amplifier 472 in accordancewith amplitude factor A_(M) generating signal s(t).

The embodiment shown in FIG. 8 differs from the other embodimentsdescribed herein in that the BPSK data is placed in the quadrature-phasechannel while the pilot data and power control data are placed in thein-phase channel. In the previous embodiments of the invention describedherein the BPSK data is placed the in-phase channel along with the pilotdata and power control data. Placing the BPSK data in thequadrature-phase channel and the pilot and power control data in thein-phase channel reduces the peak-to-average power ratio of the reverselink signal the phases of the channels are orthogonal causing themagnitude of the sum of the two channels to vary less in response tochanging data. This reduces the peak power required to maintain a givenaverage power, and thus reduces the peak-to-average power ratiocharacteristic of the reverse link signal. This reduction in thepeak-to-average power ratio decreases the peak power at which a reverselink signal must be received at the base station in order to sustain agiven transmission rate, and therefore increases the distance asubscriber unit having a maximum transmit power may be located from thebase station before it is unable to transmit a signal that can receivedat base station with the necessary peak power. This increases the rangeat which the subscriber unit can successfully conduct communication atany given data rate, or alternatively allows greater data rates to besustained at a given distance.

FIG. 9 is a block diagram of finger demodulator 182 when configured inaccordance with the embodiment of the invention shown in FIG. 8. Receivesamples R_(I) and R_(Q) are time adjusted by timing adjust 290 and thePN_(I) and PN_(Q) codes are multiplied by long code 200 usingmultipliers 301. The time adjusted receive samples are then multipliedby the complex conjugate of the PN_(I) and PN_(Q) codes usingmultipliers 302 and summers 304 yielding terms X_(I) and X_(Q). A firstand second instance of the X_(I) and X_(Q) terms are demodulated usingWalsh code W₁, and Walsh code W₂ using multipliers 310 and the resultingdemodulation symbols are summed in sets of four using summers 312. Athird instance of the X_(I) and X_(Q) terms are summed over fourdemodulation symbols by summers 308 to generate pilot reference data.The pilot reference data is filtered by pilot filters 314 and used tophase rotate and scale the summed Walsh code modulated data usingmultipliers 316 and adders 320 producing BPSK soft decision data, andafter being summed over 384 symbols by 384:1 summer 322, soft decisionpower control data.

FIG. 10 is a block diagram of a transmit system configured in accordancewith still another embodiment of the invention. Channel gain 400 gainadjusts pilot channel 402 based on gain variable A₀. Fundamental channelsymbols 404 are mapped into +1 and −1 values by mapper 405, and eachsymbol is modulated with Walsh code W_(F) equal to +,+,−,−(where +=+1and −=−1). The W_(F) modulated data is gain adjusted based on gainvariable A₁ by gain adjust 406. The outputs of gain adjusts 400 and 406are summed by summer 408 yielding in-phase data 410.

Supplemental channel symbols 411 are mapped to + and − values by mapper412, and each symbol is modulated with a Walsh code W_(S) equal to +,−.Gain adjust 414 adjusts the gain of the W_(S) modulated data. Controlchannel data 415 is mapped to + and − values by mapper 416. Each symbolis modulated with a Walsh code W_(C) equal to +, +, +, +, −, −,−, −. TheW_(C) modulated symbols are gain adjusted by gain adjust 418 based ongain variable A₃, and the output of gain adjusts 414 and 418 are summedby summer 419 to produce quadrature phase data 420.

It should be apparent that, since the Walsh codes W_(F) and W_(S) aredifferent lengths, and are generated at the same chip rate, thefundamental channel transmits data symbols at a rate that is half thatof the supplemental channel. For similar reasons, it should be apparentthat the control channel transmits data symbols at half the rate of thefundamental channel.

In-phase data 410 and quadrature phase data 420 are complex multipliedby the PN_(I) and PN_(Q) spreading codes as shown, yielding in-phaseterm X_(I) and quadrature phase term X_(Q). The quadrature phase termX_(Q) is delay by ½ the duration of a PN spreading code chip to performoffset QPSK spreading, and then term X_(I) and term X_(Q) areupconverted in accordance the R_(F) processing system 106 shown in FIG.4, and described above.

By using Walsh codes W_(F), W_(S) and W_(C) having different lengths asdescribed above, this alternative embodiment of the invention provides aset of communication channels having a greater variety of rates.Additionally, the use of a shorter, two chip, Walsh code W_(S) for thesupplemental channel provides an orthogonal higher data ratesupplemental channel with a peak-to-average transmit power ratio that isless than that associated with the use of two channels based on 4 chipWalsh codes. This further enhances the performance of the transmitsystem in that a given amplifier will be able to sustain higher rate, ortransmit with greater range, using the lower peak-to-average transmitpower waveform.

The Walsh code allocation scheme described with regard to FIG. 10, canalso be viewed as the allocation of eight chip Walsh space in accordancewith Table VI.

TABLE VI Eight-Chip Walsh Code Channel + + + +  + + + + Pilot + − + −  +− + − Supplemental + + − −  + + − − Fundamental + − − +  + − − +Supplemental + + + +  − − − − Control + − + −  − + − + Supplemental + +− −  − − + + Fundamental + − − +  − + + − Supplemental

In addition to reducing the peak to average transmit power ratio,allocating sets of eight-chip Walsh channels using a single shorterWalsh code decreases the complexity of the transmit system. For example,modulating with four eight-chip Walsh codes and summing the resultsrequires additional circuitry and therefore would be more complex.

It is further contemplated that the transmission system shown in FIG. 10can operate at various spreading bandwidths, and therefore with theWalsh codes and spreading codes generated at various rates other than1.2288 Mchips/second. In particular, a spreading bandwidth of 3.6864 MHzis contemplated, with a corresponding Walsh and spreading code rate of3.6864 Mchips/second. FIGS. 11-14 illustrate the coding performed forthe fundamental, supplemental and control channels in accordance withthe use of a 3.6864 MHz spreading bandwidth. Typically, to adjust thecoding for use with a 1.2288 MHz spreading bandwidth the number ofsymbol repeats is reduced. This principal or adjusting the number ofsymbol repeats can be applied more generally to increases in thespreading bandwidth including, for example, the use of a 5 MHz spreadingbandwidth. Adjustments performed to the coding for a 1.2288 MHzspreading bandwidth system other than reduction in the number of symbolrepeats are particularly noted in the description of FIGS. 11-14provided below.

FIG. 11 shows the coding performed for the four rates (i.e. full, half,quarter and eighth rate) that make up the IS-95 rate set 1 whenperformed in accordance with one embodiment of the invention. Data issupplied in 20 ms frames having the number of bits shown for each rate,and CRC check bits and eight tail bits are added by CRC checks sumgenerators 500 a-d and tail bit generators 502 a-d. Additionally, rate ¼convolutional encoding is performed for each rate by convolutionalencoders 504 a-d, generating four code symbols for each data bit, CRCbit, or tail bit. The resulting frame of code symbols is blockinterleaved using block interleavers 506 a-d, generating the number ofsymbols indicated. For the lower three rates, the symbols aretransmitted repeatedly by transmission repeaters 508 a-c, as indicated,causing 768 code symbols to be generated for each frame. The 768 codesymbols for each rate are then repeated 24 times by symbol repeaters 510a-d generating 18,432 code symbols per frame for each rate.

As discussed above, each code symbol in the fundamental channel ismodulated with a four bit Walsh code W_(F) generated at 3,686,400 chipsper second (3.6864 Mchips/second). Thus, for a 20 ms time interval({fraction (1/50)}th of a second) the number of Walsh and spreading codechips is 73,728, which corresponds to 4 Walsh chips for each of the18,432 code symbol in the frame.

For a system operating at 1.2288 Mchips/second, the number of symbolrepeats performed by symbol repeaters 510 a-d is reduced to eight (8).Additionally, transmission repeater 508 b repeats the sequence ofsymbols in the frame three (3) times, and a subset of the sequence ofsymbols having 120 symbols is repeated a fourth time. Transmissionrepeater 508 c repeats the sequence of symbols in the frame six (6)times, and a subset of the sequence of symbols having 48 symbols isrepeated a seventh time. Additionally, a fourth transmission repeater(or fourth transmission repeat step) is included for the full rate (notshown) which transmits 384 of the sequence of symbols contained in theframe a second time. These repeated transmissions all provide 768symbols of data which, when repeated eight times by symbol repeaters 510a-d, correspond to 6,144 symbols, which is the number of chips in a 20ms frame at 1.2288 Mchips/second.

FIG. 12 shows the coding performed for the four rates that make up IS-95rate set 2 when performed in accordance with one embodiment of theinvention. Data is supplied in 20 ms frames having the number of bitsshown for each rate, and a reserve bit is added by reserve bitaugmenters 521 a-d for each rate. CRC check bits and eight tail bits arealso added by CRC checks sum generators 520 a-d and tail bit generators522 a-d. Additionally, rate ¼ convolutional encoding is performed foreach rate by convolutional encoders 524 a-d, generating four codesymbols for each data, CRC or tail bit. The resulting frame of codesymbols is block interleaved using block interleaves 526 a-d generatingthe number of symbols indicated. For the lower three rates, the symbolsare transmitted repeatedly by transmission repeaters 528 a-c asindicated, causing 768 code symbols to be generated for each frame. The768 code symbols for each rate are then repeated 24 times by symbolrepeaters 530 a-d generating 18,432 code symbols per frame for eachrate.

For a system operating at 1.2288 MHz spreading bandwidth, the number ofsymbol repeats performed by symbol repeaters 530 a-d is reduced to four(4). Additionally, transmission repeater 528 a transmits the sequence ofsymbols in the frame two (2) times, plus 384 of the symbols aretransmitted a third time. Transmission repeater 528 b repeats thesequence of symbols in the frame five (5) times, plus 96 of the symbolsare transmitted a sixth time. Transmission repeater 528 c repeats thesequence of symbols in the frame ten (10) times, plus 96 of the symbolsare repeated an eleventh time. Additionally, a fourth transmissionrepeater (or fourth transmission repeat step) is included for the fullrate (not shown) which transmits 384 of the sequence of symbolscontained in the frame a second time. These repeated transmissions allprovide 1,536 symbols of data which, when repeated four times by symbolrepeaters 530 a-d, correspond to 6,144 symbols.

FIG. 13 illustrates the coding performed for the supplemental channelwhen performed in accordance with one embodiment of the invention.Frames of data are supplied at any of the eleven rates indicated, andCRC check sum generator 540 adds 16 bits of CRC checksum data. Tail bitgenerator 542 adds eight bits of encoder tail data resulting in frameshaving the data rates shown. Convolution encoder 544 performs rate ¼,constraint length K=9, encoding generating four code symbols four eachdata, CRC or tail bit received, and block interleaver 546 performs blockinterleaving on each frame, and outputs the number of code symbols shownfor each frame in accordance with the input frame size. Symbol repeater548 repeats the frames N times depending on the input frame size asindicated.

The encoding for an additional twelfth rate is shown, which is performedin similar fashion to the eleven rates, with the exception that rate ½encoding is performed instead of rate ¼. Additionally, no symbolrepetition is performed.

A list of frame sizes, encoder input rates, code rates and symbolrepetition factors N for various chip rates that can be applied to FIG.13 to adjust for different chip rates (which correspond to spreadingbandwidths) is provided in Table VII.

TABLE VII Encoder Symbol Chip Number Input Repetition Rate of OctetsRate Code Factor (Mcps) per Frame (kbps) Rate (N) 1.2288 21 9.6 ¼ 161.2288 45 19.2 ¼ 8 1.2288 93 38.4 ¼ 4 1.2288 189 76.8 ¼ 2 1.2288 381153.6 ¼ 1 1.2288 765 307.2 ½ 1 3.6864 21 9.6 ¼ 48 3.6864 33 14.4 ¼ 323.6864 45 19.2 ¼ 24 3.6864 69 28.8 ¼ 16 3.6864 93 38.4 ¼ 12 3.6864 14157.6 ¼ 8 3.6864 189 76.8 ¼ 6 3.6864 285 115.2 ¼ 4 3.6864 381 153.6 ¼ 33.6864 573 230.4 ¼ 2 3.6864 1,149 460.8 ¼ 1 3.6864 2,301 921.6 ½ 17.3728 21 9.6 ¼ 96 7.3728 33 14.4 ¼ 64 7.3728 45 19.2 ¼ 48 7.3728 6928.8 ¼ 32 7.3728 93 38.4 ¼ 24 7.3728 141 57.6 ¼ 16 7.3728 189 76.8 ¼ 127.3728 285 115.2 ¼ 8 7.3728 381 153.6 ¼ 6 7.3728 573 230.4 ¼ 4 7.3728765 307.2 ¼ 3 7.3728 1,149 460.8 ¼ 2 7.3728 2,301 921.6 ¼ 1 7.3728 4,6051,843.2 ½ 1 14.7456 21 9.6 ¼ 192 14.7456 33 14.4 ¼ 128 14.7456 45 19.2 ¼96 14.7456 69 28.8 ¼ 64 14.7456 93 38.4 ¼ 48 14.7456 141 57.6 ¼ 3214.7456 189 76.8 ¼ 24 14.7456 285 115.2 ¼ 16 14.7456 381 153.6 ¼ 1214.7456 573 230.4 ¼ 8 14.7456 765 307.2 ¼ 6 14.7456 1,149 460.8 ¼ 414.7456 1,533 614.4 ¼ 3 14.7456 2,301 921.6 ¼ 2 14.7456 4,605 1,843.2 ¼1 14.7456 9,213 3,686.4 ½ 1

FIG. 14 is a block diagram of the processing performed for the controlchannel for a 3.6864 MHz spreading bandwidth system. The processing issubstantially similar to that associated with the other channels, exceptfor the addition a mux 560 and symbol repeater 562, which operate tointroduce uncoded power control bits into the code symbol stream. Thepower control bits are generated at a rate of 16 per frame, and repeated18 times by symbol repeater 562 resulting in 288 power control bits perframe. The 288 power control bits are multiplexed into the frame of codesymbols at a ratio of three power control bits per coded data symbol,generating 384 total symbols per frame. Symbol repeater 564 repeats the384 bits 24 times generating 9,216 symbols per frame for an effectivedata rate of 500 kbits/second for the control data, and 800 kbits/secondfor the power control bits. The preferred processing performed for a1.2288 MHz bandwidth system simply reduces the number of symbolrepetitions performed from 24 to 8.

Thus, a multi-channel, high rate, CDMA wireless communication system hasbeen described. The description is provided to enable any person skilledin the art to make or use the present invention. The variousmodifications to these embodiments will be readily apparent to thoseskilled in the art, and the generic principles defined herein may beapplied to other embodiments without the use of the inventive faculty.Thus, the present invention is not intended to be limited to theembodiments shown herein but is to be accorded the widest scopeconsistent with the principles and novel features disclosed herein.

What is claimed is:
 1. A method of transmitting a variable data ratesignal comprising: interleaving a frame of code symbols to produce asequence of interleaved symbols having a first predetermined number ofsymbols; repeating the sequence of interleaved symbols a number oftimes; and repeating a subset of the sequence of interleaved symbols,wherein the subset has a second predetermined number of symbols, andwherein the second predetermined number of symbols is less than thefirst predetermined number of symbols.
 2. The method of claim 1 whereinthe first predetermined number of symbols is
 216. 3. The method of claim2 wherein the second predetermined number is
 120. 4. The method of claim1 wherein the first predetermined number of symbols is
 120. 5. Themethod of claim 4 wherein the second predetermined number is
 48. 6. Themethod of claim 1 wherein the second predetermined number of symbols is120.
 7. The method of claim 6 wherein the first predetermined number is216, and wherein the number of times is three.
 8. The method of claim 1wherein the second predetermined number of symbols is
 48. 9. The methodof claim 8 wherein the first predetermined number is 120, and whereinthe number of times is six.
 10. The method of claim 1 wherein the numberof times is based on a number of code symbols in the frame of codesymbols.
 11. The method of claim 1 wherein the second predeterminednumber of symbols is based on the number of code symbols in the frame ofcode symbols.
 12. A transmitter apparatus comprising: an interleaverconfigured to interleave a frame of code symbols to produce a sequenceof interleaved symbols having a first predetermined number of symbols;and a repeater configured to repeat the sequence of interleaved symbolsa number of times and to repeat a subset of the sequence of interleavedsymbols, wherein the subset has a second predetermined number ofsymbols, and wherein the second predetermined number of symbols is lessthan the first predetermined number of symbols.
 13. The apparatus ofclaim 12 wherein the first predetermined number of symbols is
 216. 14.The apparatus of claim 13 wherein the second predetermined number is120.
 15. The apparatus of claim 12 wherein the first predeterminednumber of symbols is
 120. 16. The apparatus of claim 15 wherein thesecond predetermined number is
 48. 17. The apparatus of claim 12 whereinthe second predetermined number of symbols is
 120. 18. The apparatus ofclaim 17 wherein the first predetermined number is 216, and wherein thenumber of times is three.
 19. The apparatus of claim 12 wherein thesecond predetermined number of symbols is
 48. 20. The apparatus of claim19 wherein the first predetermined number is 120, and wherein the numberof times is six.
 21. The apparatus of claim 12 wherein the number oftimes is based on a number of code symbols in the frame of code symbols.22. The apparatus of claim 12 wherein the second predetermined number ofsymbols is based on the number of code symbols in the frame of codesymbols.
 23. A transmitter apparatus comprising: means for interleavinga frame of code symbols to produce a sequence of interleaved symbolshaving a first predetermined number of symbols; and means for repeatingthe sequence of interleaved symbols at least once and for additionallyrepeating a subset of the sequence of interleaved symbols, wherein thesubset has a second predetermined number of symbols, and wherein thesecond predetermined number of symbols is less than the firstpredetermined number of symbols.